Power efficiency is a principal objective in the design of battery-powered wireless communications devices. In these types of devices the radio frequency RF output stage of the transmitter, in particular the RF power amplifier (or RF PA), is the major consumer of power. For this reason substantial efforts have been directed towards ways of improving the efficiency of the RF PA output stage in battery-powered wireless communication devices.
One approach to improving power efficiency is to employ what is known as a “switch-mode” PA in the transmitter output stage. Switch-mode PAs employ one or more switches (typically bipolar junction transistors (BJTs) or field-effect transistors (FETs)) to drive the transmitter load. As explained in more detail below, this results in a much more efficient use of power, compared to non-switched amplifier approaches.
FIG. 1 is a block diagram of a conventional switch-mode PA circuit 100, as described in U.S. Pat. No. 3,919,656 to Sokal et al. An input signal is coupled to an input terminal 102 of a driver stage 104 of the circuit 100. The driver stage 104 is configured to control an active device 108 (e.g., a BJT or FET) via a signal coupled over lead 106. The active device 108 acts substantially as a switch when appropriately driven by the driver stage 104. The output port of the active device 108 is therefore symbolically represented as a single-pole single-throw switch 110 in the drawing. Connected across the switch 110 is a series combination of a direct current (DC) powers supply 112 (e.g. a battery) and the input port of a load network 114. The output port of the load network 114 is connected to a load 116. As the switch 110 is cyclically operated at the desired alternating current (AC) output frequency, DC energy from the power supply 112 is converted into AC energy at the switching frequency.
The high efficiency attribute of switch-mode PAs derives from the fact that the transistors are operated so that they dissipate very little power. FIGS. 2A and 2B are signal waveforms, of the voltage v(t) dropped across the switch 110 and the current i(t) passed through the switch 110 of the switch-mode PA circuit 100 in FIG. 1. The drive signals applied to the switch 110 are such that the switch 110 is either in a compressed state or a cut-off state. During times when the switch 110 is compressed (i.e., when the switch is closed or ‘ON’) appreciable current flows through the switch 110, while the voltage across it is very nearly zero. During times when the switch 110 is cut-off (i.e., when the switch is open or ‘OFF’) nearly all of the voltage supplied by the power supply 112 is dropped across the switch 110, and the current flowing through it is very nearly zero. When operated in this manner the amplifier's output depends on the amplitude of the power applied to the drain (or collector if a BJT is used) of the transistor, and not on the magnitude of the signal applied to the gate (or base, in the case of a BJT) of the transistor. Because the current passed through the switch, and the voltage dropped across the switch, are never both large at the same time, very little power is dissipated by the switch.
The superior power efficiency properties of switch-mode PAs is the impetus for their use in modern battery powered RF transmitters. FIG. 3 shows, for example, how a switch-mode RF PA 316 is employed in a polar transmitter 300. The polar transmitter 300 comprises a symbol generator 302; a rectangular-to-polar converter 304; an envelope path including an envelope digital to analog converter (DAC) 306 and an envelope modulator 308; a phase path including a phase DAC 310, a phase-locked loop (PLL) including a phase modulator 312 and a voltage controlled oscillator (VCO) 314; an RF PA 316; and an antenna 318.
The polar transmitter 300 operates by first receiving a digital message to be transmitted from the symbol generator 302. Using the digital data in the digital message, the symbol generator 302 generates in-phase (I) and quadrature phase (Q) baseband signals. The I and Q baseband signals are coupled to the rectangular-to-polar converter 304, which, as the name suggests, converts the I and Q baseband signals into amplitude (i.e., ‘envelope’) and phase component signals, as indicated by the ‘ρ’ and ‘θ’ symbols in FIG. 3, respectively.
In the transmitter's phase path, the phase DAC 310 operates to convert the phase component signal into an analog waveform, which is then coupled to the phase modulator 312 and VCO 314. Based on the phase information contained in the phase component signal, the phase modulator 312 and VCO 314 then generate a phase-modulated RF carrier signal (i.e., ‘PM’ signal). Meanwhile, in the envelope path, the envelope DAC 306 operates to convert the envelope component signal (i.e., the amplitude modulation or ‘AM’ signal) into an analog waveform. This analog envelope component signal is coupled to the envelope modulator 308, which operates to modulate a power supply voltage, Vsupply (e.g., as provided by the wireless communication device's battery), according to variations in amplitude of the envelope signal. In this manner an amplitude modulated power supply signal containing the envelope information of the original input signal is created.
To generate the final modulated RF carrier signal which the antenna 318 can radiate over the air, the amplitude modulated power supply signal, VS, from the envelope path is coupled to a power supply port of the RF PA 316 while the RF PM signal from the VCO 314 in the phase path is coupled to an RF input of the RF PA 316. For a given gate (i.e. drive) voltage (assuming a FET is used for the transistor switch), as the modulated power supply voltage applied to the power supply port of the RF PA 316 is changed, the drain current of the transistor changes. This so-called ‘drain modulation’ operates to superimpose the envelope information from the envelope path onto the RF phase-modulated signal applied to the RF input of the RF PA 316 in the phase path. Because the peak amplitude of the signal into the RF PA 316 remains constant over time, linearity concerns involving amplifying non-constant envelope signals are avoided.
While use of switch-mode PAs in polar transmitters does result in a more efficient transmitter compared to more conventional quadrature modulator approaches, use of switch-mode PAs does present various problems. A first problem relates to the drive signals used to control the switching transistor of the RF PA. The drive signals applied to the switching transistor in state-of-the-art RF transmitters are sinusoidal in nature. However, sinusoidal waveforms have finite rise and fall times, which means that there are times between the ON and OFF states (described above) when the transistor switch is neither totally ON nor totally OFF. During these times the current-voltage product rises and, consequently, the power dissipated by the switching transistor also undesirably rises.
Another problem with using a switching transistor in a switch-mode PA is that the switching transistor can undesirably leak some of the drive signal applied to its input to its output. This leakage path is a well-known problem and is attributable to a parasitic capacitance formed between the input and output of the transistor (e.g., between the gate and drain of a FET type transistor or the base and collector of a BJT). To maximize efficiency the switching transistor is configured to operate in compression, a state in which the output is essentially independent of the magnitude of the signal applied to the transistor's control input. However, due to the presence of the parasitic capacitance, some of the input leaks through the parasitic capacitance to the transistor's output. This leaked signal is highly undesirable since it can cause distortion in the final RF output signal of the switch-mode PA.
One approach that might be used to avoid the power dissipation problem would be to simply increase the amplitude of the drive signals. As illustrated in FIG. 4, the increased amplitudes would have the effect of reducing the times needed to turn the transistor ON and OFF (t′ON and t′OFF, respectively) compared to the times needed without the amplification (tON and tOFF). Unfortunately, such an approach would exacerbate the leakage problem. It would also be wasteful from a power consumption perspective, since the resulting signal would have amplitudes higher than necessary to turn the transistor switch ON and OFF. In other words, excess power would be required and dissipated by the RF PA driver in order to generate the increased amplitude signals. For these reasons, simply increasing the amplitude of the sinusoidal drive signals is not, in most circumstances, a suitable solution.
Given the foregoing problems and limitations of RF PAs, it would be desirable to have methods and apparatus for reducing and/or controlling both the leakage and power dissipation of transistors used in RF PAs.